A power amplifier (PA) is a device for receiving an input signal and generating an output signal that is proportional to the input signal but has a higher amplitude. The output signal thus has a higher power than the input signal. A PA may also be referred to simply as an amplifier. The power ratio or the amplitude ratio of the output signal with respect to the input signal is known as the gain, or more specifically, as the power gain and the amplitude gain, respectively. An ideal amplifier has a gain that is independent of the power of the input signal. In practice, however, an amplifier is not able to output an arbitrarily high power. The gain of an amplifier therefore tends to become smaller with increasing input power. The power domain of the input signal for which the gain may be considered constant is known as the linear domain or linear regime. When the power of the input signal (input power) exceeds the linear domain, the amplifier is said to be saturated.
A Doherty amplifier is an amplifier that comprises a main amplifier and a peak amplifier, which are combined at input and output in parallel by, e.g., transmission lines or their lumped element equivalents. For example, the input terminal of the main amplifier may be connected to the input terminal of the peak amplifier by a transmission line having an electrical length of 90 degrees. Similarly, the output terminal of the peak amplifier may be connected to the output terminal of the main amplifier by a transmission line having an electrical length of 90 degrees. When the input power is in the linear domain of the main amplifier, the peak amplifier is switched off to save energy. The linear domain may be below an average power level of the input modulated signal. When the input power exceeds the linear domain, however, the peak amplifier is switched on to assist the main amplifier in generating the output signal. The Doherty amplifier therefore has a linear domain that is larger than both the linear domain of the main amplifier and the linear domain of the peak amplifier. The output of the main amplifier and the output of the peak amplifier may be combined in a combining node to generate the output signal of the Doherty amplifier.
An example of a conventional Doherty PA is described by making reference to FIG. 7. Ztr_in and Ztr_out are impedance transformers for matching of power devices in a required frequency band. They may have a required transformation ratio of up to 250 times for Ztr_in and above fifty times for Ztr_out, for example, to match, e.g., a 100 Watt (W) power device to standard 50 Ohm. A significant phase shift of internal matching structures inside the high power RF device, which may generally be greater than 70°, may lead to a situation where the resulting impedance transformer Ztr_out may have an overall electrical length much greater than 90°, and Ztr_out cannot be used as a Doherty combiner, too. Therefore, in order to provide a proper load modulation effect at the output of the main amplifier, the designer of such a Doherty amplifier may use a 180 degrees matching structure. Such a high transformation ratio and large electrical length of resulting Ztr_out together with an electrical length of 50 Ohm impedance inverter, e.g., 90°/Zo, as in FIG. 7, may limit the relative operational frequency band of the Doherty amplifier to about five to seven percent for Doherty amplifiers with an output power Pout greater than, e.g., 120 W.
An additional limitation of the bandwidth may be caused by dynamic load line modulation at the output of the main device, which may also introduce power dependant phase distortions (AM-PM). These may be considered as non-linear effects, and they may require additional supply energy, expensive Digital-Pre-Distortions (DPD) equipment and software to linearize the Doherty performance to the <−60 dBc level.
The circuitry for combining the output signals of the main amplifier and the peak amplifier is a 90° transmission line or lumped element equivalent having a specific characteristic impedance, referred to as a Doherty combiner. Due to demand of Mobile Communication industry for wideband, low cost, small size and highly efficient amplifiers providing excellent repeatability of parameters, the lumped element equivalents of transmission lines may be more suitable for the integration of Doherty PA. Another reason is that low-pass prototype equivalents of a transmission line may provide additional and very important advantages over a distributed line, such as:
1) a much wider range of the available characteristic impedance Zo of, e.g. 1 to 150 Ohms, at very compact physical dimensions, which also on top of that may depend very little on the operational frequency band, and;
2) harmonic rejection in the output signal of the device, allowing for a better linearity of PA.
Examples of such lumped element Doherty combiners are described in US patent publication U.S. Pat. No. 7,078,976 B2 and US patent application publication US 2010/0026387 A1 ('387), for example. Document '387 notably suggests applying a phase shift of only 90° to the main amplifier output signal before the main amplifier output signal and the peak amplifier output signal are combined, using a lumped element line. This may allow to realize the largest bandwidth of Doherty amplifier operation. A bandwidth greater than 30% may thus be achieved, at least in theory. Document '387 also suggests that distributed transmission lines may be very limited in the range of realizable Zo, which may be in the range of about 12 to 100 Ohms. When Zo is less than, e.g., 10 Ohms, the aspect ratio, i.e., the ratio length to width, of distributed transmission line on a printed circuit board (PCB) may be too small, which may give rise to electromagnetic (EM) waves with higher propagation modes in the transmission line.
Traditional high power Doherty amplifiers use, e.g., RF semiconductor power devices, which may comprise a package with input and output terminals, so called leads, and active semiconductor dies with internal matching structures or impedance transformers arranged inside the package. The package may provide mechanical strength, protection from the environment and heat dissipation, which may be up to, e.g., 200% of the maximal output RF power level of the device.
There are several issues related to the traditional packaged devices and traditional Doherty amplifier architecture are limiting an instant operational RF bandwidth, which is a parameter of great demand from modern mobile communication systems like, e.g., 4G/Long Term Evolution (LTE).
A first issue is related to internal design of traditional power devices, where the active dies are connected to the output leads by bond wires, creating parasitic inductance. This inductance may prevent direct access to the active die output or even an internal channel, the so called “voltage controlled current source”. According to theory, direct access to the current source of the power device may allow for the largest bandwidth of Doherty amplification, as proposed in U.S. Pat. No. 7,078,976 B2, where a lumped element Doherty combiner is connected directly to the current source of a power device die inside the package. A parasitic inductance between the active die and the device output terminal or lead may therefore degrade the performance of the Doherty amplifier, because it may be impossible to remove this inductance or to include it as part of the traditional Doherty combiner approach.
Secondly, when a very high power RF device, e.g., having a power greater than >200 W, several active dies of a large periphery, e.g., Wg≧100 mm, may be arranged inside a standard package with properly designed and integrated power distribution and matching networks at the input and the output of every die. If power distribution along the internal structure of the power device is not uniform, the device may become unstable and exhibit poor linearity, and lack ruggedness and reliability. Additionally, requirements for Doherty amplification friendliness may need to be fulfilled, because the designer of the Doherty amplifier may be unable to change the internal device structure to better fit for the required performance.
As mentioned above, the Doherty operational frequency bandwidth may further be affected by a power dependant phase characteristic and group delay of the main amplifier, which may be caused by a modulation of the impedance transformation at the output of the main device. This can be a fundamental issue as it is caused by the principal mechanism of the Doherty amplification concept. The group delay τ may be defined as the derivative of the phase φ characteristic versus the angular frequency ω:
      τ    ⁡          (      ω      )        =            ∂      φ              ∂      ω      
According to circuit theory, the quality of impedance matching may be expressed through a reflection coefficient Γ so, impedance transformation is related to phase of the signal as in Equation 1, where ZL is a power dependant load impedance to which an impedance transformer or inverter is connected. The larger the Zo/ZL ratio, the less bandwidth of effective matching is available according to (2). The effect is similar to the effect of quality factor Q of a resonance LC network on the phase characteristic and bandwidth. As a result, the Q-factor, amplitude and phase frequency response of the network behind the main amplifier may be affected by the variable power dependant impedance transformation ratio, caused by the signal delivered by the peak device to the common load or to the combining point of the Doherty amplifier.
                              |          Γ          |                =                  cos          ⁢                                          ⁢          θ          ⁢                                                    Z                L                            -                              Z                0                                                    2              ⁢                                                                    Z                    L                                    ⁢                                      Z                    0                                                                                                          (        1        )                                                      Δ            ⁢                                                  ⁢            f                                f            0                          =                  2          -                                    4              π                        ⁢                                          cos                                  -                  1                                            ⁡                              [                                                                            Γ                      m                                                                                      1                        -                                                  Γ                          m                          2                                                                                                      ·                                                            2                      ⁢                                                                                                    Z                            0                                                    ⁢                                                      Z                            L                                                                                                                                      |                                                                        Z                          L                                                -                                                  Z                          0                                                                    |                                                                      ]                                                                        (        2        )            
The impedance transformation ratio of the output network and thus the operational bandwidth of the main amplifier and its phase characteristic frequency response become power dependant due to a load modulation effect at the output of main device according to Equation 2. A simulation of this effect is presented in FIGS. 8, 9 and 10. It can be observed that the higher the Z transformation is (see FIGS. 8, 9, and 10), the more distortion of phase and group delay characteristic may be observed in the required frequency band. So, due to operation with a variable transformation ratio of the output network of main amplifier and the existing variable impedance mismatch in it, the electromagnetic wave of the amplified signal may move forward and be reflected backward in some portion expressed by (1). When combined at the output of the amplifier or power device represented by a voltage controlled current source, at each frequency of operational bandwidth the phase and amplitude of these two signals, i.e., the forward and the reflected signals may depend in the Doherty amplifier also on the instant power level, which may affect linearity of the phase characteristic of the amplifier for each current power level. The resulting AM-PM characteristic of the Doherty amplifier or device may thus become different at different frequencies of Doherty operation. But uniformity of AM-PM versus frequency of operation is one of the primary parameter of state-of-the-art Doherty PAs, as it may limit the operational bandwidth and linearization capability. According to latest requirements for power amplifiers operating in 4G/LTE mobile base stations, their phase characteristic deviation from the linear one in operational bandwidth must be less than 2°, otherwise linearization of such a power amplifier with a Digital Pre-Distortion (DPD) system may be problematic.
FIG. 8 shows a graph of the reflection coefficient as a function of the phase shift of the signal passing through the impedance transformer Ztr_out at the output of the main amplifier for different transformation ratios in the range of 1 to 50 connected as shown in FIG. 7.
FIG. 9 shows a graph of the same reflection coefficient as a function of the relative frequency deviation of the signal passing through the impedance transformer for different transformation ratios in the range of 1 to 50 as a demonstration of a bandwidth limiting effect of output matching Ztr_out of main device.
FIG. 10 illustrates an example of phase and group delay frequency response non-linear distortions (“unwrap(phase(S(1,2)))” and “S.delay(2,1)) ” for impedance transformation ratios of 1, 2, 3, and 5 (thick line), respectively introduced by the impedance inverter, or by the Doherty combiner, at the output terminal of the main amplifier, which may occur along power variation delivered by the peak amplifier according to the basic mechanism of the Doherty amplification principle. It is a clear frequency band degradation with an increase of the impedance modulation ratio.
At 10% of the frequency offset from the center of bandwidth (see FIG. 10), at a transformation ratio of 3:1, a phase deviation from a linear response may achieve about 7°. It means that the AM-PM amplitude of the Doherty amplifier made of, e.g., ideal distributed transmission lines, may be expected to be about 7° larger or, depending on the Doherty architecture, smaller on the side than in the center of the frequency band. In contrast, a mobile BTS multi-carrier PA may be required to have a phase characteristic that deviates by less than 2° from linear in the operational bandwidth.
The time delay Tg of the signal passing through the network having resonance at a frequency fo and a quality factor Q may be estimated as
                    Tg        -                                            Q                                                                          π                ⁢                                                                  ⁢                fo                                                                        (        3        )            
An ideal distributed transmission line impedance inverter as a Doherty combiner (90°/Zo in FIG. 7) introduces no amplitude or phase response distortions at the peak power level of the Doherty amplifier, where the source and the load impedances are equal to the characteristic impedance of the impedance inverter as Zs=Zl=Zo (e.g., ZO=50 Ohms), where the quality factor Q of this system is equal to 1. At the peak power level, phase and amplitude response distortions of the Doherty amplifier may thus be introduced by the impedance transformer Ztr_out, because Doherty combiner is matched to 50 Ohms at both sides, while at back-off power level below average its contribution to the Doherty amplifier non-linearity of phase and amplitude response versus frequency is maximum. When an impedance variation occurs at Zl of the Doherty amplifier, and also at the output of the main amplifier, caused by the signal delivered by the peak device in the range between the average power level and the peak power level of Doherty, then Q≠1 and Zo≠Zs≠Zl. The quality factor Q of the main amplifier output network may thus become a function of the power delivered by the peak device and may vary about 2 times for a symmetrical Doherty amplifier and up to, e.g., >3 times for an asymmetrical 2:1 Doherty configuration. An ideal impedance inverter may be represented by and behave more like a resonator with some effective variable quality factor Q, e.g., as presented in FIG. 10, which may define the frequency bandwidth of the signal which may pass through the output network of the main amplifier to the Doherty load without significant distortions. Accordingly, the time delay of the signal may become variable with respect to both the power of the peak device and the Q factor as expressed by formula (3). As a result of this effect, the wideband signal which passes through the output network of the main device to the load may suffer from time propagation distortions, and the signal arriving at the load of the Doherty amplifier is not the same as the signal delivered by the output of the main device, distortion of which may depend on the power level or amplitude of the amplified signal. If presented in the time domain, this signal may already have a different shape, which will be different for a different power level, and not because of power saturation of active device, but because of different propagation times for different frequencies of the band. To the best knowledge of the inventors, this effect which may occur in a Doherty amplifier has not been described before.
The solution presented below addresses the Doherty amplifier bandwidth. It may include: 1) improving the matching configuration of the Doherty output combiner and 2) minimizing the power dependency of phase characteristic and group delay of the main and peak amplifiers.
FIG. 11 illustrates the simplest correcting LCR network example and FIG. 12 the group delay and phase characteristics in the band for the impedance inverter and corrector.
FIG. 12 shows, by way of an example, the characteristics of distortions of phase, cf. “unwrap(phase(S(1,2)))” and delay, cf. S.delay(2,1), in the frequency band created by an ideal inverter as a Doherty combiner at 3:1 load modulation (bold lines) and also correction characteristics (thin lines of an exemplary parallel LCR, as a corrector circuit, on the same plot. To correct properly, they may exhibit opposite trends but the same value at every power level. When such a LCR corrector is connected in series to the input of the main amplifier, it may be able to correct phase distortions in the back-off power range below the average power level at high impedance of the main device load built of ideal transmission lines. When the power level of the Doherty amplifier exceeds the average power P, the value of the resistance R in the LCR may be expected to drop to R=0 at peak power level, where the correction effect of the corrector will be minimal or close to zero. If R is replaced, for example, by a linear voltage dependent resistor, or by a channel of a field effect transistor (FET) controlled by, e.g., an envelope signal, the group delay distortion can be corrected by an envelope or so called base-band modulation signal of a very low power level, which will not affect the overall power efficiency of such a Doherty amplifier.
A similar phase or group delay distortion correction effect may be achieved by a series RLC network, e.g., as a dual to the one described above, connected in parallel to the input of the main amplifier, where the FET can be connected as a series or parallel component to the RLC, providing the required dynamic Q-factor and phase characteristic deviation.
To control the R value, a radio frequency (RF) envelope detector may be required, which can be used for R value control along the power level.
FIG. 13 shows a Doherty amplifier architecture that has been modified accordingly, allowing a dynamic linearization effect for phase frequency response and AM-PM distortions over the required wide frequency band. The figure includes a Zcorr line of a certain electrical length at the input of the peak amplifier, to compensate for the additional delay which may be introduced by the RLC corrector at the input of the main amplifier.
It should be mentioned here that in case of lumped element equivalent matching structures which may be used for the purpose of integration at the output of the main amplifier instead of distributed transmission lines and also different matching configuration using, for example, a high-pass or band-pass filter prototype instead of, e.g., a low-pass prototype configuration, the frequency response and the impedance transformation properties may be different, so the correcting network required at the input of the main device may be different too. Expression (4) represents the relation between the group delay Tg and the cut-off frequency Fc of the Low-Pass filter of the prototype network and the same for a band-pass network (5) having bandwidth BW:
                    Tg        =                  1                      4            ⁢            π            ⁢                                                  ⁢            Fc                                              (        4        )                                Tg        =                  1                      2            ⁢            π            ⁢                                                  ⁢            BW                                              (        5        )            